Electric motor controller

ABSTRACT

An electric motor controller includes a first coordinate conversion unit that converts a three-phase current into a current in a δc-γc rotating coordinate system, a first calculation unit that obtains a first term, a second calculation unit that obtains a voltage command value in the δc-γc rotating coordinate system as a sum of the first term and a second term, a second coordinate conversion unit that coordinate-converts the voltage command value into a voltage command value of a voltage to be applied to a rotary electric motor in another coordinate system.

TECHNICAL FIELD

The present technique relates to controlling a synchronous motorcomprising a field and an armature.

More particularly, the present technique relates to controlling a rotaryelectric motor on the basis of a so-called primary magnetic flux whichis a synthesis of a field flux that the field generates and a magneticflux of an armature reaction generated by an armature current flowing inthe armature.

BACKGROUND ART

Conventionally, various controls of a rotary electric motor on the basisof a primary magnetic flux, i.e., a so-called primary magnetic fluxcontrols have been proposed. Briefly speaking, the primary magnetic fluxcontrol is a technique for stably controlling the rotary electric motorby controlling the primary magnetic flux of the rotary electric motor inaccordance with a command value thereof.

It is assumed, for example, that a phase of a field flux Λ0 is employedat a d axis in rotating coordinate system, a phase of a primary magneticflux λ1 is employed at a δ axis in another rotating coordinate system,and a phase difference of the δ axis with respect to the d axis is aload angle φ. Herein, however, a γ axis is employed at a 90-degreeleading phase with respect to the δ axis. Further, a δc axis and a γcaxis are defined as control axes in the rotating coordinate system whichis employed in the primary magnetic flux control. The δc axis and the γcaxis are corresponding to the δ axis and the γ axis, respectively, and aphase difference of the δc axis with respect to the d axis is assumed asφc.

In this case, a command value of the primary magnetic flux λ1(hereinafter, referred to as a “primary magnetic flux command value”)has a δc-axis component Λδ*, and a γc-axis component is zero. Therefore,when the primary magnetic flux λ1 is equal to the primary magnetic fluxcommand value, the δc-axis component λ1δc of the primary magnetic fluxλ1 is equal to the δc-axis component Λδ*, the phase difference φc isequal to the load angle φ, and the δc axis is coincident with the δaxis.

The δc-axis component λ1δc and the γc-axis component λ1γc of the primarymagnetic flux λ1 vary with a change of the primary magnetic flux commandvalue, a variation in a load, an influence of control disturbance,or/and the like. For example, the change of the primary magnetic fluxcommand value and the variation in the load invites a transient changeof the primary magnetic flux λ1, and the control disturbance invites avariation in the γc axis/δc axis. As states where the controldisturbance occurs, for example, a state where a voltage applied to therotary electric motor is different from a voltage command due to aninfluence of a time delay, an on-loss, and dead time, and a state wherethere is a deviation between a device constant of the rotary electricmotor and that assumed by a control system. Therefore, a deviationarises between the primary magnetic flux λ1 and the primary magneticflux command value, and accordingly a deviation also arises between theload angle φ and the phase difference φc.

In the primary magnetic flux control, when there is a deviation betweenthe primary magnetic flux λ1 and the primary magnetic flux commandvalue, a control, for example, of a voltage command value to becorrected is performed so that the δc-axis component λ1δc of the primarymagnetic flux λ1 may be made equal to the δc-axis component Λδ* of theprimary magnetic flux command value and the γc-axis component λ1γc ofthe primary magnetic flux λ1 may become zero. The phase difference φc isthereby coincident with the load angle φ.

In such a primary magnetic flux control, control is made with a torqueof the rotary electric motor being made in direct proportion to aγc-axis component of an armature current, not depending on a rotationangular velocity thereof.

Among the following prior-art documents, in Yabe and Sakanobe, “ASensor-less Drive of IPM Motor with Over-modulation PWM”, The papers ofJoint Technical Meeting on Rotating Machinery, IEE Japan, 2001 (159),pp. 7 to 12, the γc axis and the δc axis are exchanged and employed, ascompared with those in the other prior-art documents.

SUMMARY Problems to be Solved

In Japanese Patent No. 3672761, a feedback is achieved by using adeviation in a δ-axis component, not by using a γ-axis component of thearmature current. Further, in Kaku, Yamamura, and Tsunehiro, “A NovelTechnique for a DC Brushless Motor Having No Position-Sensors”, IEEJTransaction on Industry Applications, 1991, Volume 111, No. 8, pp. 639to 644, assumed is a range in which the load angle φ can equallyapproximate a sine value sirup thereof.

In any one of the above documents, however, except Yabe and Sakanobe, “ASensor-less Drive of IPM Motor with Over-modulation PWM”, The papers ofJoint Technical Meeting on Rotating Machinery, IEE Japan, 2001 (159),pp. 7 to 12, with respect to an inductance of the armature winding, ad-axis component thereof and a γ-axis component which is 90-degree phaseadvance therewith are handled isotropically, and the technique cannot beapplied to a rotary electric motor having so-called saliency such as aninterior permanent magnet rotary electric motor.

Further, the amount of feedbacks employed in any one of the prior-artdocuments does not include any information of the load angle φ. Forexample, a δ-axis current and a γ-axis current are employed in JapanesePatent Application Laid Open Gazette No. 4-91693 and Hotta, Asano, andTsunehiro, “Method of controlling Position Sensorless DC brushlessmotor”, 1988 Tokai-Section Joint Conference of the Institutes ofElectrical and Related Engineers, p. 161, Kaku and Tsunehiro, “A NovelTechnique for a DC Brushless Motor Having No Position-Sensors”, 1990Tokai-Section Joint Conference of the Institutes of Electrical andEngineers, p. 172, Kaku, Yamamura, and Tsunehiro, “A Novel Technique fora DC Brushless Motor Having No Position-Sensors”, IEEJ Transaction onIndustry Applications, 1991, Volume 111, No. 8, pp. 639 to 644, and Yabeand Sakanobe, “A Sensor-less Drive of IPM Motor with Over-modulationPWM”, The papers of Joint Technical Meeting on Rotating Machinery, IEEJapan, 2001 (159), pp. 7 to 12 and the δ-axis current is employed inJapanese Patent No. 3672761 and Urita, Tsukamoto, and Tsunehiro,“Constant estimation method for synchronous machines with the primarymagnetic flux controlled”, 1998 Tokai-Section Joint Conference of theInstitutes of Electrical Engineers, p. 101 and Urita, Yamamura, andTsunehiro, “On General Purpose Inverter for Synchronous Motor Drive”,IEEJ Transaction on Industry Applications, 1999, Volume 119, No. 5, pp.707 to 712, respectively, for the amount of feedbacks. For this reason,in an area where the load angle φ is large, the primary magnetic fluxcannot be controlled to a desired value. When a large torque isoutputted, the load angle φ also becomes large. Therefore, in theconventional primary magnetic flux control, it is hard to appropriatelyperform a stable drive or a high-efficient drive in the area where thetorque is large.

In order to solve the above problem, a technique is described to applythe primary magnetic flux control to the rotary electric motor evenhaving saliency by performing a feedback based on the deviation of theprimary magnetic flux. Also described is a technique to provide aprimary magnetic flux control in which a drive can be performed at astable and high-efficient operating point even in an area where anoutput torque is large.

Means for Solving the Problems

An electric motor controller according to the present disclosure is adevice for controlling a primary magnetic flux ([λ1]) on a rotaryelectric motor including an armature having an armature winding and arotor which is a field rotating relatively to the armature, the primarymagnetic flux being a synthesis of a field flux (Λ0) that the fieldgenerates and a magnetic flux (λa: id·Ld, iq·Lq) of an armature reactiongenerated by an armature current ([I]) flowing in the armature.

A first aspect of the electric motor controller according to the presentdisclosure includes a first coordinate conversion unit (101) thatconverts the armature current into a first current ([i]) in a rotatingcoordinate system (δc-γc) having a predetermined phase (φc) with respectto rotation of the rotor, a first calculation unit (102) that sums aninductive voltage (ω*·[Λ1*]) by a primary magnetic flux command value([Λ1*]) which is a command value of the primary magnetic flux and avoltage drop ({R}[i]) by the first current on the basis of a voltageequation at a time when the rotary electric motor to obtain a first term([F]), a second calculation unit (103A, 103B) that sums said first termand a second term ([B]) obtained by performing an operation expressed bya non-zero matrix ({K}) on a deviation ([ΔΛ]) of the primary magneticflux with respect to the primary magnetic flux command value to obtain afirst voltage command value ([v*]) which is a command value of a voltageto be applied to the rotary electric motor in the rotating coordinatesystem, and a second coordinate conversion unit (104) thatcoordinate-converts the first voltage command value into a secondvoltage command value ([V*]) which is a command value of the voltage tobe applied to the rotary electric motor in another coordinate system.

A second aspect of the electric motor controller according to thepresent disclosure is the first aspect thereof in which the secondcalculation unit (103A) employs an estimation value ([λ1̂]) of theprimary magnetic flux as the primary magnetic flux.

A third aspect of the electric motor controller according to the presentdisclosure, which is the second aspect thereof, further includes aprimary magnetic flux estimation unit (105) that obtains the estimationvalue ([λ1̂]) of the primary magnetic flux from the predetermined phase(φc), a first component (Lq) orthogonal to the field flux of aninductance of the armature winding, a second component (Ld) in phasewith the field flux of the inductance, the first current, and the fieldflux (Λ0).

A fourth aspect of the electric motor controller according to thepresent disclosure, which is the second or third aspect thereof, furtherincludes a primary magnetic flux command correction unit (107) thatcorrects the primary magnetic flux command value ([Λ1*]) to output aprimary magnetic flux command correction value ([Λ1**]) by using thepredetermined phase (φc), a first component (Lq) orthogonal to the fieldflux of an inductance of the armature winding, a second component (Ld)in phase with the field flux of the inductance, the first current, thefield flux (Λ0), and the estimation value ([λ1̂]) of the primary magneticflux. The second calculation unit (103B) employs the primary magneticflux command correction value as the primary magnetic flux commandvalue.

An estimation value of the predetermined phase may be employed as thepredetermined phase. For example, the predetermined phase (φc) isobtained from the first voltage command value ([v*]), a resistance value({R}) of the armature winding, the first component (Lq), a rotationangular velocity (ω*) of the rotor, and the first current ([i]).

Effects

In the electric motor controller of the first aspect according to thepresent disclosure, since the second term obtained on the basis of thedeviation of the primary magnetic flux functions as a feedback for thefirst voltage command value, the second term has information of a loadangle, and even when the deviation between the predetermined phase andthe load angle is large, it becomes easier to perform the primarymagnetic flux control while correcting the deviation. Further, theprimary magnetic flux control does not depend on whether or not there issaliency.

In the electric motor controller of the second aspect according to thepresent disclosure, it is not necessary to perform a direct detection ofthe primary magnetic flux.

In the electric motor controller of the third aspect according to thepresent disclosure, it is possible to perform the primary magnetic fluxcontrol while correcting the deviation of the load angle regardless ofwhether or not there is saliency.

In the electric motor controller of the fourth aspect according to thepresent disclosure, it is possible to achieve accuracy on the same levelwith the second or third aspect, regardless of a method of detecting orestimating the primary magnetic flux.

These and other objects, features, aspects and advantages of the presentdisclosure will become more apparent from the following detaileddescription of the present disclosure when taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF DRAWINGS

FIGS. 1 to 3 are vector diagrams illustrating a first preferredembodiment;

FIGS. 4 and 5 are block diagrams illustrating the first preferredembodiment;

FIG. 6 is a block diagram illustrating a second preferred embodiment;

FIG. 7 is a vector diagram illustrating the second preferred embodiment;

FIG. 8 is a block diagram illustrating the second preferred embodiment;

FIGS. 9 and 10 are block diagrams illustrating a third preferredembodiment; and

FIG. 11 is a vector diagram illustrating a variation.

DESCRIPTION OF EMBODIMENTS

In the following embodiments, description will be made with athree-phase interior permanent magnet rotary electric motor taken anexample. It is obvious that a rotary electric motor of polyphase, otherthan three-phase, as well as a rotary electric motor other than aninterior permanent magnet type is also applicable.

The First Preferred Embodiment

FIGS. 1 and 2 are vector diagrams each illustrating a primary magneticflux control.

In the primary magnetic flux control, set is a δc-γc coordinate systemwhich is in phase advance with a d-q coordinate system (a d axis inphase with a field flux Λ0, a q axis is in 90-degree phase advance withthe d axis) with reference to a phase of the field flux Λ0 (i.e. withthe rotation of a rotor), by a phase difference φc. Then, a voltage tobe applied to the rotary electric motor (a γc-axis component and aδc-axis component thereof are assumed to be vγc and vδc, respectively)is adjusted so that a δc axis may be coincident with a δ axis, which isin phase with the primary magnetic flux.

First, FIG. 1 shows a case where the phase difference φc is coincidentwith the load angle φ. As shown in FIG. 1, a magnetic flux λa of anarmature reaction is a synthesis of a magnetic flux Lq·iq in a q-axispositive direction and a magnetic flux Ld·id in a d-axis negativedirection. Then, the primary magnetic flux is a synthesis of themagnetic flux λa and the field flux Λ0 and takes a positive value Λδ(coincident with a command value Λδ* thereof) in the δ axis (coincidentwith the δc axis in FIG. 1).

An inductive voltage ω·Λδ(=ω·Λδ*) by the primary magnetic flux appearson a γc axis (herein, coincident with a γ axis). Further, forexplanatory convenience, when it is grasped that the inductive voltageω·Λ0 in a case where the armature reaction is disregarded (in otherwords, it is assumed that the magnetic flux λa=0) is an inductivevoltage by the field flux, the inductive voltage ω·Λ0 appears on the qaxis.

Therefore, the inductive voltage by the armature reaction is representedas a synthesis of a voltage ω·Lq·iq in the d-axis negative direction anda voltage ω·Ld·id in a q-axis negative direction.

Introducing a resistance value R of an armature winding, a voltage dropby an armature current appears as a voltage R·iδc on the δc axis and asa voltage R·iγc on the γc axis.

Therefore, assuming that the γc-axis component and the δc-axis componentof the voltage to be applied to the rotary electric motor are a voltagevγc and a voltage vδc, respectively, when the primary magnetic flux iscoincident with the primary magnetic flux command value,vγc−R·iγc=ω·Λδ*, vδc=R·iδc are true, as shown in FIG. 1.

Now, the δc-axis component λ1δc and/or the γc-axis component λ1γc of theprimary magnetic flux λ1 vary with a variation in the load, an influenceof control disturbance, or/and the like. Therefore, as shown in FIG. 2,a deviation arises between the phase difference φc and the load angle φ.From the definition of the δ axis, since the primary magnetic has noγ-axis component, the primary magnetic flux which actually occurs isalso referred to as a primary magnetic flux Λδ.

In the δc-γc rotating coordinate system on which the primary magneticflux control is performed, a control is made so that the δc-axiscomponent λ1δc of the primary magnetic flux Λδ may be coincident with (aδc-axis component Λδ* of) the primary magnetic flux command value andthe γc-axis component λ1γc of the primary magnetic flux Λδ may becoincident with (a γc-axis component Λγ*=0 of) the primary magnetic fluxcommand value.

In order to make the δc-axis component λ1δc coincident with the δc-axiscomponent Λδ* of the primary magnetic flux command value, the inductivevoltage ω·Λδ* on the γc axis needs to appear. Also in consideration ofthe voltage drop in the armature winding, it is necessary to set thevoltage command value as a sum of the inductive voltage ω·Λδ* and thevoltage drop. Herein, the sum is represented as a feedforward term[F]=[FγFδ]^(t) (the former component represents the γc-axis componentand the latter component represents the δc-axis component: thesuperscript “t” represents a transpose of a matrix: the same applies tothe following unless otherwise indicated). Eqs. (1) and (2) are derivedfrom a voltage equation of a rotary electric motor, where a differentialoperator p is introduced.

$\begin{matrix}{\lbrack F\rbrack = {{\left\{ R \right\} \lbrack i\rbrack} + {\begin{Bmatrix}p & \omega^{*} \\{- \omega^{*}} & p\end{Bmatrix}\left\lbrack {{\Lambda 1}*} \right\rbrack}}} & (1) \\{\begin{bmatrix}{F\; \gamma} \\{F\; \delta}\end{bmatrix} = {{\begin{Bmatrix}R & 0 \\0 & R\end{Bmatrix}\begin{bmatrix}i & \gamma & c \\i & \delta & c\end{bmatrix}} + \begin{bmatrix}{{\omega^{*} \cdot \Lambda}\; \delta^{*}} \\0\end{bmatrix} + \begin{bmatrix}0 \\{{p \cdot \Lambda}\; \delta^{*}}\end{bmatrix}}} & (2)\end{matrix}$

In Eq. (1), it can be grasped that a matrix {R} is a tensor indicating aresistance of the armature winding, and as shown in Eq. (2), the matrix{R} has the same component R both on the δc axis and the γc axis andoff-diagonal components are zeros. Further, a current vector [i]=[iγciδc]^(t) indicating a current flowing the armature winding isintroduced. The first term on the right side of each of Eqs. (1) and (2)represents a voltage drop {R}[i]. The third term of Eq. (2) is atransient term and can be disregarded. This is because an influence ofthe transient term can be also handled as the deviation between the loadangle φ and the phase difference φc, as described above.

Further, assuming herein that both the δc axis and the δ axis rotatewith respect to the d axis at an angular velocity ω which is equal to acommand value ω* of the angular velocity, ω=ω*. By appropriatelyperforming the primary magnetic flux control, ω=ω* is true.

When the phase difference φc is equal to the load angle φ, since theδc-axis component Λδ* of the primary magnetic flux command value[Λ1*]=[0 Λδ*]^(t) in the δc-γc rotating coordinate system is coincidentwith the primary magnetic flux Λδ, the feedforward term [F] is thevoltage command value [v*] for the rotary electric motor (also see FIG.1).

When the γ axis is not coincident with the γc axis, however, the phase(φc−φ) is not resolved only by employing only the feedforward term [F]as the voltage command value. In the primary magnetic flux control,since no control is made on the basis of the deviation with respect tothe voltage command value [v*] of the voltage [v] to be applied to therotary electric motor, a voltage deviation [ve]=[v]−[v*] arises. Thephase difference (φc−φ) thereby remains. Therefore, in order to resolvethe phase difference (φc−φ) (in order to make φc=φ), as the voltagecommand value [v*]=[vγc* vδc*]^(t) to be determined with respect to theprimary magnetic flux Λδ, a vector represented at a position obtained byrotationally transfer the feedforward term [F] represented at a positionJ1 to be in phase advance by the phase difference (φc−φ) (in thecounterclockwise direction in FIG. 2) should be employed. This isbecause only with the feedforward term [F], a voltage [v] arises at aposition in phase lag with the voltage command value [v*] by the phasedifference (φc−φ).

The matrix operation of such rotational transfer of the vector, however,cannot be performed. This is because the load angle φ which actuallyarises is not known.

As is clear from FIG. 2, the difference between the positions J1 and J2is caused by the difference [ΔΛ]=[0−λ1γc Λδ*−λ1δc]^(t) between themagnetic flux in the primary magnetic flux control, which takes theprimary magnetic flux command value Λδ* in the δc-γc rotating coordinatesystem (on the δc axis), and the primary magnetic flux Λδ which actuallyarises in a δ-γ rotating coordinate system (on the δ axis). Thedifference is grasped, by the substances thereof, as the deviation ofthe primary magnetic flux with respect to the primary magnetic fluxcommand value.

Therefore, by calculating the voltage command value [v*] with the sum ofthe feedback term [B]=[Bγ Bδ]^(t) and the feedforward term [F] (see FIG.3), in spite of the presence of the voltage deviation [ve], it ispossible to reduce the difference between the feedforward term [F] andthe voltage [v]. Now, the feedback term [B], however, can be obtained byEqs. (3) and (4).

$\begin{matrix}{\lbrack B\rbrack = {\left\{ K \right\} \lbrack{\Delta\Lambda}\rbrack}} & (3) \\{\begin{bmatrix}{B\; \gamma} \\{B\; \delta}\end{bmatrix} = {\begin{Bmatrix}{K\; {\gamma\gamma}} & {K\; {\gamma\delta}} \\{K\; {\delta\gamma}} & {K\; \delta \; \delta}\end{Bmatrix}\begin{bmatrix}{{- \gamma}\; 1\; \gamma \; c} \\{{\Lambda\delta}^{*} - {\gamma \; 1\; \delta \; c}}\end{bmatrix}}} & (4)\end{matrix}$

At least one of components Kγγ, Kγδ, Kδγ, and Kδδ in a matrix {K} forperforming an arithmetic operation on the deviation [ΔΛ] of the magneticflux is not zero. In other words, the matrix {K} is a non-zero matrix.

The feedforward term [F] functions as a feedforward based on thearmature current and the feedback term [B] functions as a feedback basedon the deviation of the magnetic flux.

When both the two elements of a column vector [Kγγ Kδγ]^(t) forming thematrix {K} are not zero, for example, the γc-axis component (−λ1γc) ofthe deviation of the magnetic flux can be fed back to the voltagecommand value [v*] with respect to both the γc axis and the δc axis.Alternatively, when both the two elements of a column vector [KγδKδδ]^(t) are not zero, the δc-axis component (Λδ*−λ1δc) of the deviationof the magnetic flux can be fed back to the voltage command value [v*]with respect to both the γc axis and the δc axis.

Further, when both the column vectors [Kγγ Kδγ]^(t) and [Kγδ Kδδ]^(t)are non-zero vectors, the magnetic flux component of both axes can befed back, and it is therefore possible to improve stability andresponsibility of the control system.

Since the feedback term [B] functions as the feedback based on thedeviation [ΔΛ] with respect to the voltage command value, if the phasedifference φc deviates with respect to the load angle φ, it becomeseasier to perform the primary magnetic flux control by correcting thedeviation. In conformity with FIG. 3, the deviation [ΔΛ] decreases, thephase difference φc approximates the load angle φ, and the γc axisapproximates the γ axis. Then, when such a feedback proceeds and the γcaxis becomes coincident with the γ axis, λ1γc=0 and λ1δc=Λδ*, and thestate shown in FIG. 1 is achieved. In other words, FIG. 3 is a vectordiagram showing a condition while the phase difference φc approximatesthe load angle φ.

As is clear from Eq. (4), the voltage command value can be determined inconsideration of the feedback based on the deviation [ΔΛ] of the primarymagnetic flux. The matrix {K} functioning as a feedback gain may or maynot have a diagonal component or an off-diagonal component only if thematrix {K} is a non-zero matrix. Further, each component may include anintegral element.

On the basis of the above idea, FIG. 4 is a block diagram showing aconfiguration of an electric motor controller 1 in accordance with thepresent embodiment and its peripheral devices.

A rotary electric motor 3 is a three-phase electric motor, and includesa not-shown armature and a rotor which is a field. As a technical commonsense, the armature has an armature winding and the rotor rotatesrelatively to the armature. The field includes, for example, a magnetwhich generates a field flux. Herein, description will be made on a casewhere an interior permanent magnet type is adopted.

A voltage supply source 2 includes, for example, a voltage control typeinverter and a control unit thereof, and applies a three-phase voltageto the rotary electric motor 3 on the basis of a three-phase voltagecommand value [V*]=[Vu* Vv* Vw*]^(t). A three-phase current [I]=[Iu IvIw]^(t) thereby flows in the rotary electric motor 3. The componentswhich the voltage command value [V*] and the three-phase current [I]have are, for example, described as a U-phase component, a V-phasecomponent, and a W-phase component in this order.

The electric motor controller 1 is a device for controlling the primarymagnetic flux [λ1] and the rotation velocity (hereinafter, illustratedas the rotation angular velocity) on the rotary electric motor 3. Theprimary magnetic flux [λ1] is a synthesis of the field flux Λ0 that afield magnet generates and the magnetic flux λa (see the components ofFIG. 1, id·Ld, iq·Lq in FIG. 1) of the armature reaction generated bythe armature current (this is also the three-phase current [I]) flowingin the armature. The magnitude of the primary magnetic flux [λ1] is acomponent Λδ on the actual δ axis, and is represented as[λ1]=[λ1γcλ1δc]^(t) in the δc-γc rotating coordinate system. In thepresent embodiment, the primary magnetic flux [λ1] is handled as anobservable value or an already-estimated value.

The electric motor controller 1 includes a first coordinate conversionunit 101, a first calculation unit 102, a second calculation unit, 103A,a second coordinate conversion unit 104, and an integrator 106.

The first coordinate conversion unit 101 converts the three-phasecurrent [I] into a current [i] in the δc-γc rotating coordinate systemwhere the primary magnetic flux control is performed.

The first calculation unit 102 obtains the feedforward term [F]. Thesecond calculation unit 103A obtains the voltage command value [v*] inthe δc-γc rotating coordinate system as a sum of the feedforward term[F] and the feedback term [B].

The second coordinate conversion unit 104 performs a coordinateconversion of the voltage command value [v*] into a voltage commandvalue [V*] of a voltage to be applied to the rotary electric motor 3 inanother coordinate system. This “another coordinate system” may be, forexample, a d-q rotating coordinate system, an α-β fixed coordinatesystem (for example, the α-axis is set in phase with the U phase), or auvw fixed coordinate system, or a polar coordinate system. Which one ofthe coordinate systems is employed as “another coordinate system”depends on which control the voltage supply source 2 performs. Forexample, when the voltage command value [V*] is set in the d-q rotatingcoordinate system, [V*]=[Vd* Vq*]^(t) (where the former component is thed-axis component and the latter component is the q-axis component).

The integrator 106 calculates a phase θ of the δc axis with respect tothe α axis on the basis of the rotation angular velocity ω. On the basisof the phase θ, the first coordinate conversion unit 101 and the secondcoordinate conversion unit 104 can perform the above coordinateconversion. The rotation angular velocity ω is obtained as an output ofa subtracter 109. The rotation angular velocity ω is obtained bysubtracting a Km times multipled value, in a constant multiplier unit108, of the γc-axis component iγc of the current [i] that has beenremoved its DC component in a high pass filter 110, from a command valueω* of the rotation angular velocity in a subtracter 109. When theprimary magnetic flux control is appropriately performed, ω=ω* asdescribed above.

FIG. 5 is a block diagram showing configurations of the firstcalculation unit 102 and the second calculation unit 103A. In FIG. 5, areference sign “x” surrounded by a circle represents a multiplier, areference sign “+” surrounded by a circle represents an adder, and acircle to which reference signs “+−” are attached represents asubtracter. Since the resistance value R, the primary magnetic fluxcommand value Λδ*=0 on the γc axis, feedback gains Kγγ, Kγδ, Kδγ, andKδδ are already known, these can be set in the first calculation unit102 and the second calculation unit 103A.

The Second Embodiment

The present embodiment shows a technique in which the electric motorcontroller 1 obtains an estimation value [λ1̂] of the primary magneticflux [λ1].

As shown in FIG. 6, the configuration of the electric motor controller 1of the present embodiment further includes a primary magnetic fluxestimation unit 105 in that of the electric motor controller 1 of thefirst embodiment. As the primary magnetic flux [λ1], the secondcalculation unit 103A employs the estimation value [λ1̂] thereof.

In general, the phase of the field flux Λ0 is employed on the d axis,and a q axis which is in 90-degree phase advance therewith is assumed.When such a d-q rotating coordinate system rotates at the angularvelocity ω, introducing a d-axis voltage vd which is a d-axis componentof the voltage to be applied to the rotary electric motor, a q-axisvoltage vq which is a q-axis component of the voltage to be applied tothe rotary electric motor, a d-axis inductance Ld which is a d-axiscomponent of the inductance of the armature winding, a q-axis inductanceLq which is a q-axis component of the inductance of the armaturewinding, and the differential operator p, Eq. (5) is held.

$\begin{matrix}{\begin{bmatrix}{vd} \\{vq}\end{bmatrix} = {{\begin{Bmatrix}{R + {p \cdot {Ld}}} & {{{- \omega} \cdot L}\; q} \\{\omega \cdot {Ld}} & {R + {p \cdot {Lq}}}\end{Bmatrix}\begin{bmatrix}{i\; d} \\{i\; q}\end{bmatrix}} + \begin{bmatrix}0 \\{\omega \cdot {\Lambda 0}}\end{bmatrix}}} & (5)\end{matrix}$

The above equation is expressed in a ξ-η rotating coordinate systemhaving a ξ axis rotating while maintaining the phase difference ψ withrespect to the d axis and a η axis in 90-degree phase advance with the ιaxis, the following Eqs. (6), (7), and (8) are held. Note thatintroduced are a ξ-axis component iξ of the armature current, a η-axiscomponent iη of the armature current, a ξ-axis component vξ and a η-axiscomponent vη of the voltage to be applied to the rotary electric motor,and a ξ-axis component λξ and a η-axis component λη of the primarymagnetic flux. Herein, it is not assumed that the primary magnetic fluxcontrol is performed.

$\begin{matrix}{\begin{bmatrix}{v\; \eta} \\{v\; \xi}\end{bmatrix} = {{\begin{Bmatrix}R & 0 \\0 & R\end{Bmatrix}\begin{bmatrix} & \eta \\i & \xi\end{bmatrix}} + {\begin{Bmatrix}p & \omega \\{- \omega} & p\end{Bmatrix}\begin{bmatrix}\lambda & \eta \\\lambda & \xi\end{bmatrix}}}} & (6) \\{\begin{bmatrix}\lambda & \xi \\{- \lambda} & \eta\end{bmatrix} = {\begin{Bmatrix}{\left( {{Lq} - {Ld}} \right)\sin \; {\psi \cdot \cos}\; \psi} & {{{{Ld} \cdot \cos^{2}}\psi} + {{{Lq} \cdot \sin^{2}}\psi}} \\{{{{- {Ld}} \cdot \sin^{2}}\psi} - {{{Lq} \cdot \cos^{2}}\psi}} & {{- \left( {{Lq} - {Ld}} \right)}\sin \; {\psi \cdot \cos}\; \psi}\end{Bmatrix}{\quad{\begin{bmatrix}i & \eta \\i & \xi\end{bmatrix} + \begin{bmatrix}{{{\Lambda 0} \cdot \cos}\; \psi} \\{{{\Lambda 0} \cdot \sin}\; \psi}\end{bmatrix}}}}} & (7) \\{{\tan \; \psi} = \frac{{v\; \xi} - {{R \cdot i}\; \xi} + {{\omega \cdot {Lq} \cdot i}\; \eta}}{{v\; \eta} - {{R \cdot i}\; \eta} - {{\omega \cdot {Lq} \cdot i}\; \xi}}} & (8)\end{matrix}$

The first term on the right side of Eq. (7) is a magnetic flux (armaturereaction) generated by the armature current flow, and the second termthereof is a magnetic flux contributing to the field flux Λ0.

Since the Eqs. (6), (7), and (8) are held regardless of the phasedifference ψ, if the phase difference ψ is replaced with the phasedifference φc, in other words, the ξ-η rotating coordinate system isreplaced with the δc-γc rotating coordinate system, the meanings of Eqs.(6), (7), and (8) are not changed. Since a phase of the actual primarymagnetic flux Λδ having the load angle φ with respect to the d axis istaken on the δ axis, with the above replacement, the value λξ representsthe δc-axis component λ1δc of the primary magnetic flux Λδ and the valueλη represents the γc-axis component λ1γc of the primary magnetic flux Λδin Eq. (7). The vector diagram at that time is shown in FIG. 7.

Therefore, from the phase difference φc, the d-axis inductance Ld, theq-axis inductance Lq, the armature currents iγc and iδc, and the fieldflux Λ0, the estimation value of the primary magnetic flux [λ1],[λ1̂]=[λ1γĉ λ1δĉ]^(t) is obtained by Eqs. (9) and (10).

$\begin{matrix}{\left\lbrack {\lambda \; 1^{\bigwedge}} \right\rbrack = {{\left\{ L \right\} \lbrack i\rbrack} + \left\lbrack {\Lambda \; 0} \right\rbrack}} & (9) \\{\begin{bmatrix}{\lambda \; 1\; \gamma \; c^{\bigwedge}} \\{\lambda \; 1\; \delta \; c^{\bigwedge}}\end{bmatrix} = {\begin{Bmatrix}{{{{Ld} \cdot \sin^{2}}{\varphi c}} + {{{Lq} \cdot \cos^{2}}\varphi \; c}} & {\left( {{Lq} - {Ld}} \right)\sin \; \varphi \; {c \cdot \cos}\; \varphi \; c} \\{\left( {{Lq} - {Ld}} \right)\sin \; \varphi \; {c \cdot \cos}\; \varphi \; c} & {{{{Ld} \cdot \cos^{2}}\varphi \; c} + {{{Lq} \cdot \sin^{2}}\varphi \; c}}\end{Bmatrix}{\quad{\begin{bmatrix}i & \gamma & c \\i & \delta & c\end{bmatrix} + \begin{bmatrix}{{{- {\Lambda 0}} \cdot \sin}\; \varphi \; c} \\{{{\Lambda 0} \cdot \cos}\; \varphi \; c}\end{bmatrix}}}}} & (10) \\{{\varphi \; c} = {\tan^{- 1}\frac{{v\; \delta \; c} - {{R \cdot i}\; \delta \; c} + {{\omega \cdot {Lq} \cdot i}\; \gamma \; c}}{{v\; \delta \; c} - {{R \cdot i}\; \delta \; c} - {{\omega \cdot {Lq} \cdot i}\; \delta \; c}}}} & (11)\end{matrix}$

Herein, introduced is a field flux vector [Λ0]=[−Λ0 sin φc Λ0·cosφc]^(t), representing the field flux Λ0 in the δc-γc rotating coordinatesystem.

Further, it can be grasped that a matrix {L} in Eq. (9) is a coefficientof the current vector [iγc iδc]^(t) of the first term on the right sidein Eq. (10) and a tensor in which the inductance of the armature windings expressed in the δc-γc rotating coordinate system. When the rotaryelectric motor has no saliency, since Ld=Lq, as is clear from FIG. 10,the off-diagonal component of the matrix {L} is zero. In other words,Eq. (10) can be employed in the rotary electric motor having saliency.

It can be grasped that the first term on the right side of each of Eqs.(9) and (10) is the magnetic flux caused by the armature reaction.

Further, the phase difference φc can employ an estimated value on thebasis of Eq. (11). In this case, the used voltages vγc and vδc mayemploy the already-obtained voltage command values vγc* and vδc* to beused for estimation of a new phase difference φc.

FIG. 8 is a block diagram showing a configuration of the primarymagnetic flux estimation unit 105. The primary magnetic flux estimationunit 105 includes a delay unit 105 a, a load angle estimating unit 105b, an armature reaction estimation unit 105 c, a field flux vectorgeneration unit 105 d, and an adder 105 e.

The armature reaction estimation unit 105 c inputs thereto the phasedifference φc, the d-axis inductance Ld, the q-axis inductance Lq, andthe armature currents iγc and iδc, and calculates the first term on theright side of Eq. (10). FIG. 8 uses the expression {L}[i] of the firstterm on the right side of Eq. (9), and that two values of the γc-axiscomponent and the δc-axis component are outputted is indicated by twoslashes.

The field flux vector generation unit 105 d inputs thereto the fieldflux Λ0 and calculates the second term on the right side of Eq. (10).FIG. 8 uses the expression [Λ0] of the second term on the right side ofEq. (9), and that two values of the γc-axis component and the δc-axiscomponent are outputted is indicated by two slashes.

The adder 105 e performs addition in the two components, the γc-axiscomponent and the δc-axis component, to thereby achieve addition of thefirst term and the second term on the right side in each of Eqs. (9) and(10), and outputs the estimation value [λ1̂] of the primary magneticflux.

In order to estimate the phase difference φc, used are the voltagecommand values vγc* and vδc* obtained by the second calculation unit103A at the immediately preceding control timing. In other words, thedelay unit 105 a delays the voltage command values vγc* and vδc*obtained by the second calculation unit 103A and the load angleestimating unit 105 b calculates the phase difference φc in accordancewith Eq. (11) at the immediately following control timing. Further,instead of employing the voltage command values vγc* and vδc* obtainedat the immediately preceding control timing, the voltage command valuesvγc* and vδc* which have been obtained at this point in time may beemployed. In this case, the delay unit 105 a may be omitted.

In the present embodiment, it is not necessary to perform directdetection of the primary magnetic flux. Further, the primary magneticflux can be estimated, regardless of whether or not there is saliency,and the primary magnetic flux control is performed while correcting thedeviation of the phase difference φc.

Thus, by performing estimation of the primary magnetic flux with thephase difference φc which is a parameter having a strong correlationwith an output torque, it is possible to estimate the primary magneticflux with high accuracy even in the area where the output torque islarge. This makes a drive of the rotary electric motor 3 stable in thearea where the output torque is large, in other words, an area where therotary electric motor 3 can be driven stably is extended. Further, evenin the area where the output torque is large, the rotary electric motor3 can be driven at a high-efficient operating point.

The Third Embodiment

In the present embodiment, shown is a technique to achieve the effectshown in the second embodiment when the electric motor controller 1obtains the estimation value or a measured value of the primary magneticflux [λ1].

As shown in FIG. 9, the electric motor controller 1 of the presentembodiment has a constitution in which the second calculation unit 103Ais replaced with a second calculation unit 103B and a primary magneticflux command correction unit 107 is further included in the constitutionof the electric motor controller 1 of the second embodiment.

Now, it is assumed that the primary magnetic flux [λ1]=[λ1γc λ1δc]^(t)is estimated by a method other than that shown in the second embodiment.A correction value [Λγ** Λδ**]^(t) of the primary magnetic flux command(hereinafter, also referred to as a primary magnetic flux commandcorrection value [Λ1**]), which satisfies the following Eq. (12)together with the primary magnetic flux [λ1], is obtained by Eq. (13).In this equation, introduced is the estimation value [λ1̂] of the primarymagnetic flux which is described in the second embodiment. Further, foreasy understanding, a γ-axis component Λγ* of the primary magnetic fluxcommand value is also clearly specified (actually, Λγ*=0).

δ*−λ1δc=

δ**−λ1δĉ

γ*−λ1γc=

γ**−λ1γĉ  (12)

∴

δ**=

δ*+λ1δĉ−λ1δc

γ**=

γ*+λ1γĉλ1γc  (13)

By performing the primary magnetic flux control in the secondembodiment, the right side of Eq. (12) becomes zero. Therefore, when theprimary magnetic flux control is performed on the primary magnetic flux[λ1] on the basis of the primary magnetic flux command correction value[Λ1**] obtained by Eq. (13), the same effect as produced in the secondpreferred embodiment can be achieved. In other words, it is natural thatit is not necessary to perform the direct detection of the primarymagnetic flux, and further, it is possible to perform the primarymagnetic flux control while correcting the deviation of the phasedifference φc, not depending on the method of measuring or estimatingthe primary magnetic flux [λ1], regardless of whether or not there issaliency.

In this case, it is not necessary to replace the primary magnetic fluxcommand value [Λ1*] in the feedforward term [F] with the primarymagnetic flux command correction value [Λ1**]. As can be understood fromFIG. 2, this is because the inductive voltage ω·Λδ* appearing on the γcaxis is determined, regardless of whether or not there is the deviation[ΔΛ].

On the other hand, the feedback term [B] is determined on the basis ofthe deviation between the primary magnetic flux [λ1] and the primarymagnetic flux command correction value [Λ1**]. Therefore, introducingthe deviation of the primary magnetic flux, [ΔΛ′]=[Λγ**−λ1γcΛδ**−λ1δc]^(t), the feedback term [B] is obtained by the followingequations.

$\begin{matrix}{\lbrack B\rbrack = {\left\{ K \right\rbrack \left\lbrack {\Delta \; \Lambda^{\prime}} \right\rbrack}} & (14) \\{\begin{bmatrix}{\beta\gamma} \\{\beta \; \delta}\end{bmatrix} = {\begin{Bmatrix}{K\; {\gamma\gamma}} & {K\; \gamma \; \delta} \\{K\; \delta \; \gamma} & {K\; {\delta\delta}}\end{Bmatrix}\begin{bmatrix}{{\Lambda\gamma}^{**} - {\lambda \; 1\gamma \; c}} \\{{\Lambda\delta}^{**} - {\lambda \; 1\; \delta \; c}}\end{bmatrix}}} & (15)\end{matrix}$

FIG. 10 is a block diagram showing a configuration of the firstcalculation unit 102 and a second calculation unit 103B. As describedabove, since the feedforward term [F] uses the primary magnetic fluxcommand value [λ1*], instead of the primary magnetic flux commandcorrection value [Λ1**], the first calculation unit 102 is employed alsoin the present embodiment, like in the first and second preferredembodiments.

On the other hand, since the calculation for obtaining the feedback term[B] uses the primary magnetic flux command correction value [Λ1**], thesecond calculation unit 103B has a configuration which is slightlydifferent from that of the second calculation unit 103A. Specifically,since Λγ*=0 in the second calculation unit 103A, this is not inputtedbut is prepared in the second calculation unit 103A. On the other hand,in the second calculation unit 103B, the γc-axis component Λγ** of theprimary magnetic flux command correction value [Λ1**] is inputted.Further, though the command value Λδ* is inputted in the secondcalculation unit 103A, the δc-axis component Λδ** of the primarymagnetic flux command correction value [Λ1**] is inputted in the secondcalculation unit 103B. In the configuration shown in FIG. 10, otherconfigurations are the same as the configuration shown in FIG. 5.

The primary magnetic flux command correction unit 107 inputs thereto theprimary magnetic flux command value [Λ1*], the estimation value [λ1̂] ofthe primary magnetic flux (calculated by the primary magnetic fluxestimation unit 105 as described in the second embodiment), and theprimary magnetic flux [λ1] which is estimated by another method. Then,by performing the calculation of Eq. (13), the primary magnetic fluxcommand correction value [Λ1**] is outputted.

<Variations>

Estimations of the primary magnetic flux [λ1] by other methods otherthan the method shown in the second embodiment will be exemplifiedbelow.

With reference to FIG. 7, in consideration of the γc-axis component(vγc−R·iγc) and the δc-axis component (vδc-R≠iδc) of an internalinductive voltage ω·Λδ, the estimation values λ1γĉ and λ1δĉ of theprimary magnetic flux Λδ are obtained as −(vδc−R iδc)/ω and(vγc−R·iγc)/ω, respectively.

Further, when the estimation value Λδ̂ of the primary magnetic flux Λδ isobtained, with reference to FIG. 7, setting χ=φ−φc, an estimation valueχ̂ of the angle χ is obtained by Eq. (16).

$\begin{matrix}{\chi^{\bigwedge} = {\tan^{- 1}\frac{- \left( {{v\; \delta \; c} - {{R \cdot i}\; \delta \; c}} \right)}{{v\; \gamma \; c} - {{R \cdot i}\; \gamma \; c}}}} & (16)\end{matrix}$

Therefore, the estimation values λ1γc ̂ and λ1δĉ are obtained as−sin(χ̂)·Λδ̂ and cos(χ̂)·Λδ̂, respectively.

Now, the estimation value Λδ̂ of the primary magnetic flux Λδ can becalculated by using, for example, the estimation value of the primarymagnetic flux in the α-β fixed coordinate system of the rotary electricmotor 3. Herein, the α-β fixed coordinate system has the α axis and theβ axis, and employs the β axis in 90-degree phase advance with the αaxis. As described earlier, for example, the α axis is adopted in phasewith the U phase.

Introducing an α-axis component λ1α̂ and a β-axis component λ1β̂ of theestimation value Λδ̂ of the primary magnetic flux Λδ, the estimationvalue Λδ̂ of the primary magnetic flux Λδ is obtained by Eq. (17).

δ̂=√{square root over (λ1α̂²+λ1β̂²)}  (17)

Now, as shown in Eq. (18), the α-axis component λ1α̂ and the β-axiscomponent λ1β̂ can be obtained by integration of the α-axis component V0αand the β-axis component V0β of the internal inductive voltage ω·Λδ withrespect to the time. The α-axis component V0α can be calculated asVα−R·iα from an a-axis component Vα of an applied voltage V observedoutside and an α-axis component iα of the current [I] flowing in therotary electric motor 3. Similarly, the β-axis component V0β can becalculated as Vβ-R·iβ from a β-axis component Vβ of the applied voltageV observed outside and a β-axis component iβ of the current [I] flowingin the rotary electric motor 3. The applied voltage V is obtained from,for example, the three-phase voltage supplied from the voltage supplysource 2 to the rotary electric motor 3 in conformity with FIG. 4.

λ1α̂=∫(

0α)dt=∫(

α−R·iα)dt

λ1β̂=∫(

0β)dt=∫(

β−R·iβ)dt  (18)

Further, when the a-axis component λ1α̂ and the β-axis component λ1β̂ areobtained, the estimation values λ1γĉ and λ1δĉ can be also obtained byanother method. In other words, the estimation values λ1γĉ and λ1δĉ canbe obtained by Eq. (19) by using the phase θ of the δc axis with respectto the α axis.

$\begin{matrix}{\begin{bmatrix}{\lambda \; 1\gamma \; c^{\bigwedge}} \\{\lambda \; 1\delta \; c^{\bigwedge}}\end{bmatrix} = {\begin{pmatrix}{{- \sin}\; \theta} & {\cos \; \theta} \\{\cos \; \theta} & {\sin \; \theta}\end{pmatrix}\begin{bmatrix}{\lambda \; 1\alpha^{\bigwedge}} \\{\lambda \; 1\beta^{\bigwedge}}\end{bmatrix}}} & (19)\end{matrix}$

Further, the α-axis component λ1α̂ and the β-axis component λ1β̂ can beobtained by another method. As described above, since the appliedvoltage V can be obtained from the three-phase voltage supplied from thevoltage supply source 2 to the rotary electric motor 3, the U-phasecomponent Vu, the V-phase component Vv, and the W-phase component Vw canbe measured. As described above, the three-phase current Iu, Iv, and Iwflowing in the rotary electric motor 3 can be measured. Therefore, theU-phase component λ1u ̂, the V-phase component λ1v̂, and the W-phasecomponent λ1ŵ of the estimation value Λδ̂ of the primary magnetic flux Λδcan be obtained by Eq. (20), like by Eq. (18).

λ1û=∫(

u−R·I u)dt

λ1v̂=∫(

v−R·I

)dt

λ1ŵ=∫(

w−R·Iw)dt  (20)

By performing the coordinate conversion of the UVW-phases and the α-βfixed coordinate system, the a-axis component λ1α̂ and the β-axiscomponent λ1β̂ can be obtained by Eq. (21). Therefore, by further usingEq. (19), the estimation values λ1γĉ and λ1δĉ can be obtained.

$\begin{matrix}{\begin{bmatrix}{\lambda \; 1\alpha^{\bigwedge}} \\{\lambda \; 1\beta^{\bigwedge}}\end{bmatrix} = {\begin{pmatrix}\sqrt{2\text{/}3} & {- \sqrt{1\text{/}6}} & {- \sqrt{1\text{/}6}} \\0 & {1\text{/}\sqrt{2}} & {{- 1}\text{/}\sqrt{2}}\end{pmatrix}\begin{bmatrix}{\lambda \; 1u^{\bigwedge}} \\{\lambda \; 1v^{\bigwedge}} \\{\lambda \; 1w^{\bigwedge}}\end{bmatrix}}} & (21)\end{matrix}$

When complete integration is performed in the integral calculation ofEqs. (18) and (20), the DC component is superimposed and the error inthe estimation of the magnetic flux thereby becomes larger. Therefore,it is preferable to perform the well-known incomplete integration.

Further, instead of Eq. (11), the phase difference φc can be estimatedas follows. Though FIG. 11 corresponds to FIG. 7, a q′ axis is newlyemployed. Herein, the q′ axis is adopted in phase with a voltage V′. Thevoltage V′ is a synthesis of the inductive voltage ω·Λδ caused by theprimary magnetic flux and a voltage having a δc-axis component ω·Ld·iγcand a γc-axis component (−ω·Ld·iδc).

Introducing a leading phase angle φc′ of the γc axis viewed from the q′axis and a leading phase angle ξ of the q′ axis viewed from the q axis,an estimation value of the phase difference φc can be obtained as a sumof the angles φc′ and ξ. Then, the angles φc′ and ξ can be obtained byEqs. (22) and (23), respectively.

$\begin{matrix}{{\varphi \; c^{\prime}} = {\tan^{- 1}\frac{{v\; \delta \; c} - {{R \cdot i}\; \delta \; c} + {{\omega \cdot {Ld} \cdot i}\; \gamma \; c}}{{v\; \gamma \; c} - {{R \cdot i}\; \gamma \; c} - {{\omega \cdot {Ld} \cdot i}\; \delta \; c}}}} & (22) \\\begin{matrix}{\xi = {\cos^{- 1}\left( {{\omega \cdot {\Lambda 0}}\text{/}{V^{\prime}}} \right)}} \\{= {\cos^{- 1}\left( \frac{{\omega \cdot \Lambda}\; 0}{\sqrt{\begin{matrix}{\left( {{v\; \gamma \; c} - {{R \cdot i}\; \gamma \; c} - {{\omega \cdot {Ld} \cdot i}\; \delta \; c}} \right)^{2} +} \\\left( {{v\; \delta \; c} - {{R \cdot i}\; \delta \; c} - {{\omega \cdot {Ld} \cdot i}\; \gamma \; c}} \right)^{2}\end{matrix}}} \right)}}\end{matrix} & (23)\end{matrix}$

In any one of the above-described preferred embodiments, the electricmotor controller 1 includes a microcomputer and a memory device. Themicrocomputer executes each process step (in other words, eachprocedure) described in a program. The above memory device can beconstituted of one or a plurality of memory devices such as a Read OnlyMemory (ROM), a Random Access Memory (RAM), a rewritable nonvolatilememory (Erasable Programmable ROM (EPROM) or the like), a hard diskunit, and the like. The memory device stores therein various informationand data and the like, also stores therein a program to be executed bythe microcomputer, and provides a work area for execution of theprogram.

It can be grasped that the microcomputer functions as various meanscorresponding to each of the process steps described in the program, orthat the microcomputer implements various functions corresponding toeach of the process steps. Further, the electric motor controller 1 isnot limited thereto, and some or all of the various procedures executedby the electric motor controller 1, or some or all of the various meansor various functions implemented by the electric motor controller 1 maybe achieved by hardware.

While the disclosure has been shown and described in detail, theforegoing description is in all aspects illustrative and notrestrictive. It is therefore understood that numerous modifications andvariations can be devised without departing from the scope of thedisclosure.

1-9. (canceled)
 10. An electric motor controller which is a device forcontrolling a primary magnetic flux on a rotary electric motor includingan armature having an armature winding and a rotor which is a fieldrotating relatively to said armature, said primary magnetic flux being asynthesis of a field flux that said field generates and a magnetic fluxof an armature reaction generated by an armature current flowing in saidarmature, comprising: a first coordinate conversion unit that convertssaid armature current into a first current in a rotating coordinatesystem having a predetermined phase with respect to rotation of saidrotor; a first calculation unit that sums an inductive voltage by aprimary magnetic flux command value which is a command value of saidprimary magnetic flux and a voltage drop by said first current on thebasis of a voltage equation of said rotary electric motor to obtain afirst term; a second calculation unit that sums said first term and asecond term obtained by performing an operation expressed by a non-zeromatrix on a deviation of said primary magnetic flux with respect to saidprimary magnetic flux command value to obtain a first voltage commandvalue which is a command value of a voltage to be applied to said rotaryelectric motor in said rotating coordinate system; and a secondcoordinate conversion unit that coordinate-converts said first voltagecommand value into a second voltage command value which is a commandvalue of said voltage to be applied to said rotary electric motor inanother coordinate system.
 11. The electric motor controller accordingto claim 10, wherein said second calculation unit employs an estimationvalue of said primary magnetic flux as said primary magnetic flux. 12.The electric motor controller according to claim 11, further comprising:a primary magnetic flux estimation unit that obtains said estimationvalue of said primary magnetic flux from said predetermined phase, afirst component orthogonal to said field flux of an inductance of saidarmature winding, a second component in phase with said field flux ofsaid inductance, said first current, and said field flux.
 13. Theelectric motor controller according to claim 11, further comprising: aprimary magnetic flux command correction unit that corrects said primarymagnetic flux command value to output a primary magnetic flux commandcorrection value by using said predetermined phase, a first componentorthogonal to said field flux of an inductance of said armature winding,a second component in phase with said field flux of said inductance,said first current, said field flux, and said estimation value of saidprimary magnetic flux, wherein said second calculation unit employs saidprimary magnetic flux command correction value as said primary magneticflux command value.
 14. The electric motor controller according to claim12, further comprising: a primary magnetic flux command correction unitthat corrects said primary magnetic flux command value to output aprimary magnetic flux command correction value by using saidpredetermined phase, said first component, said second component, saidfirst current, said field flux, and said estimation value of saidprimary magnetic flux, wherein said second calculation unit employs saidprimary magnetic flux command correction value as said primary magneticflux command value.
 15. The electric motor controller according to claim12, wherein an estimation value of said predetermined phase is employedas said predetermined phase.
 16. The electric motor controller accordingto claim 13, wherein an estimation value of said predetermined phase isemployed as said predetermined phase.
 17. The electric motor controlleraccording to claim 14, wherein an estimation value of said predeterminedphase is employed as said predetermined phase.
 18. The electric motorcontroller according to claim 15, wherein said estimation value of saidpredetermined phase is obtained from said primary magnetic flux commandvalue, a resistance value of said armature winding, said firstcomponent, a rotation angular velocity of said rotor, and said firstcurrent.
 19. The electric motor controller according to claim 16,wherein said estimation value of said predetermined phase is obtainedfrom said primary magnetic flux command value, a resistance value ofsaid armature winding, said first component, a rotation angular velocityof said rotor, and said first current.
 20. The electric motor controlleraccording to claim 17, wherein said estimation value of saidpredetermined phase is obtained from said primary magnetic flux commandvalue, a resistance value of said armature winding, said firstcomponent, a rotation angular velocity of said rotor, and said firstcurrent.